1. Field of the Invention
The present invention relates to a receiver system. More particularly, the present invention relates to a receiver system provided with a filter circuit employing an operational transconductance amplifier.
2. Description of the Prior Art
A receiver system is usually provided with a filter circuit in the form of an integrated circuit. When a filter circuit including an inductor is formed into an integrated circuit, since the inductor is difficult to integrate, it is customary to use, instead of an inductor having one end grounded as shown in FIG. 13A, an equivalent inductor circuit L1 as shown in FIG. 13B and, instead of a floating inductor as shown in FIG. 14A, an equivalent inductor circuit L2 as shown in FIG. 14B.
The equivalent inductor circuit L1 of FIG. 13B is composed of operational transconductance amplifiers (hereinafter referred to as OTAs) 1 and 2 and a capacitor C1. The output terminal of the OTA 1 and the non-inverting input terminal of the OTA 2 are connected together, and the node between these serves as an end of the equivalent inductor circuit L1. The inverting input terminal of the OTA 1 and the output terminal of the OTA 2 are connected together, and the node between these is connected to one end of the capacitor C1. The other end of the capacitor C1, the non-inverting input terminal of the OTA 1, and the inverting input terminal of the OTA 2 are grounded. The equivalent inductance L1 of the equivalent inductor circuit L1 is given by formula (1) below, where C1 represents the reactance of the capacitor C1, and gm represents the conductance of each of the OTAs 1 and 2.L1=C1/(gm)2  (1)
On the other hand, the equivalent inductor circuit L2 of FIG. 14B is composed of OTAs 3, 4, and 5, and a capacitor C2. The output terminal of the OTA 3 and the non-inverting input terminal of the OTA 4 are connected together, and the node between these serves as one end of the equivalent inductor circuit L2. The inverting input terminal of the OTA 4 and the output terminal of the OTA 5 are connected together, and the node between these serves as the other end of the equivalent inductor circuit L2. The inverting input terminal of the OTA 3, the output terminal of the OTA 4, and the non-inverting input terminal of the OTA 5 are connected together, and the node among these is connected to one end of the capacitor C2. The other end of the capacitor C2, the non-inverting input terminal of the OTA 3, and the inverting input terminal of the OTA 5 are grounded. The equivalent inductance L2 of the equivalent inductor circuit L2 is given by formula (2) below, where C2 represents the reactance of the capacitor C2, and gm represents the conductance of each of the OTAs 3, 4, and 5.L2=C2/(gm)2  (2)
Ideally, an equivalent inductor circuit is equivalent to an inductor having no resistance; in reality, however, it includes resistance. As an example, a Smith chart in FIG. 15 shows the impedance characteristics of the equivalent inductor circuit L1 where C1=3.7 [pF] and gm=165 [μS].
The imaginary part of the impedance of the equivalent inductor circuit L1 becomes greater as the frequency of the input signal becomes higher. Since the imaginary part of the impedance of the equivalent inductor circuit L1 remains positive irrespective of the frequency of the input signal, the equivalent inductor circuit L1 functions as an inductor.
On the other hand, the real part of the impedance of the equivalent inductor circuit L1 becomes smaller as the frequency of the input signal becomes higher, and eventually becomes negative when the frequency of the input signal becomes higher than 900 kHz. That is, the impedance of the equivalent inductor circuit L1 comes to include negative resistance when the frequency of the input signal becomes higher than 900 kHz.
The presence of such negative resistance leads to oscillation. The impedance characteristics of the equivalent inductor circuit L2 are similar to those of the equivalent inductor circuit L1.
When a filter circuit is formed into an integrated circuit, a resistor having one end grounded as shown in FIG. 16A is often replaced with an equivalent resistor circuit R1 as shown in FIG. 16B. The equivalent resistor circuit R1 of FIG. 16B is composed of an OTA 6. The output terminal and the inverting input terminal of the OTA 6 are connected together, and the node between these serves as an end of the equivalent resistor circuit R1. The non-inverting input terminal of the OTA 6 is grounded. The equivalent resistance R1 of the equivalent resistor circuit R1 is given by formula (3) below, where gm represents the conductance of the OTA 6.R1=1/gm  (3)
FIG. 17 shows the configuration of a band-pass filter circuit, as an example of a conventional filter circuit employing the equivalent inductor and resistor circuits described above.
An input terminal 7 is connected to one end of an equivalent inductor circuit L3. The other end of the equivalent inductor circuit L3 is connected to one end of a capacitor C3. The other end of the capacitor C3 is connected to one end of a capacitor C4, to an equivalent inductor circuit L4, and to one end of an equivalent inductor circuit L5. The other end of the capacitor C4 is grounded, and the other end of the equivalent inductor circuit L5 is connected to one end of a capacitor C5.
The other end of the capacitor C5 is connected to one end of a capacitor C6, to an equivalent inductor circuit L6, to an equivalent resistor circuit R2, and to an output terminal 8. The other end of the capacitor C6 is grounded.
Here, the equivalent inductor circuits L3 and L5 have the same configuration as the equivalent inductor circuit L2 shown in FIG. 14B, and the equivalent inductor circuits L4 and L6 have the same configuration as the equivalent inductor circuit L1 shown in FIG. 13B. The equivalent resistor circuit R2 has the same configuration as the equivalent resistor circuit R1 shown in FIG. 16B.
When the circuit constants of the band-pass filter circuit of FIG. 17 are so set that ƒC=2 MHz, the gain characteristics obtained exhibit, as shown in FIG. 18, undesirable peaks near the lower cutoff frequency ƒC1 and the upper cutoff frequency ƒC2. This results from the above-described impedance characteristics of the equivalent inductor circuits, specifically, the presence of negative resistance in the impedance of the equivalent inductor circuits L3 to L6 in the frequency band above 900 kHz. A receiver system, when provided with a band-pass filter circuit with such inadequate gain characteristics, does not offer satisfactory reception performance.
Moreover, in the band-pass filter circuit of FIG. 17, the constants of the individual circuit elements are determined arbitrarily, and the different circuit elements have different individual variations originating from their fabrication. This makes it impossible to reduce variations in the cutoff frequencies, which are determined by those circuit constants. To obtain the cutoff frequencies as designed, a band-pass filter circuit is sometimes so configured as to be adjustment-free by being provided with a phase control loop. However, even in this configuration, the equivalent inductor circuits provided in the filter circuit (for example, a low-pass filter circuit) provided in the phase control loop and those provided in the band-pass filter circuit have negative resistance. Thus, the individual filter circuits have unsatisfactory gain characteristics, and produce great errors in the actually obtained cutoff frequencies from their design values.
Incidentally, one type of receiver system is superheterodyne receiver apparatuses. In a superheterodyne receiver apparatus, a band-pass filter is provided in the stage following a mixer that down-converts a received RF (radio-frequency) signal and outputs an IF (intermediate-frequency) signal. The band-pass filter serves to eliminate unnecessary frequency components from the IF signal.
In superheterodyne receiver apparatuses that handle IF signals in a frequency band of from about 1 to 3 MHz, a band-pass filter for eliminating unnecessary frequency components from the IF signal is generally built as a band-pass filter circuit (hereinafter referred to also as a gm band-pass filter) employing operational transconductance amplifiers as shown in FIG. 17 and described above. This permits the integration of the band-pass filter for eliminating unnecessary frequency components from the IF signal.
On the other hand, in superheterodyne receiver apparatuses that handle IF signals in a frequency band of from about 100 to 200 MHz, it is necessary to use a band-pass filter of a high order to eliminate unnecessary frequency components from the IF signal. Accordingly, here, the band-pass filter for eliminating unnecessary frequency components from the IF signal is generally built not as a gm band-pass filter but as a SAW (surface-acoustic-wave) filter or the like.
The gm band-pass filter of FIG. 17 has the inductors L3 to L6 built as equivalent inductor circuits employing operational transconductance amplifiers, and thus can be integrated. However, the gm band-pass filter of FIG. 17 includes active elements (transistors) inside the operational transconductance amplifiers, and thus suffers from distortion in the input-output characteristics. This distortion causes intermodulation.
One commonly used indicator of the degree of distortion is the third-order input intercept point. Now, with reference to FIG. 19, which shows the distortion characteristics of the gm band-pass filter of FIG. 17, the third-order input intercept point will be explained. The third-order intercept point IP3′ is the intersection point between the extension line of the linear portion of the curve representing the output 107 of the target signal (the signal at the center frequency of the gm band-pass filter of FIG. 17) with respect to the input signal and the extension line of the linear portion of the curve representing the output 108 of the third-order intermodulation distortion with respect to the input signal. The third-order input intercept point IIP3′ represents the level of the input signal at the third-order intercept point IP3′.
Here, the output 108 of the third-order intermodulation distortion is determined by feeding two signals, having frequencies of 5 MHz and 8 MHz respectively and having identical levels, to the gm band-pass filter of FIG. 17 and measuring the levels of the third-order intermodulation distortion appearing in the output signal, i.e., the levels of a 2 (2×5−8) MHz signal and a 11 (2×8−5) MHz (this method is called two-tone measurement).
The higher the third-order input intercept point IIP3′, the less the gm band-pass filter of FIG. 17 is affected by interfering waves. With the gm band-pass filter of FIG. 17, however, the third-order input intercept point IIP3′ is too low, specifically, −2 dBm. Moreover, here, the third-order input intercept point IIP3′ is not expected to be improved by the adjustment of the circuit constants. A receiver system, when provided with a gm band-pass filter with too low a value of the third-order input intercept point IIP3′, does not offer satisfactory reception performance.